1. Field of the Invention
The present invention generally relates to radars, and more particularly to a DSSS (Direct Sequence Spectrum Spreading) radar, a method implemented by a DSSS radar, and a computer-readable storage medium which stores a program for causing a computer to carry out such a method.
2. Description of the Related Art
Generally, the radar is also referred to as a radiolocater, and detects the position and velocity of a target by irradiating a radio wave with respect to the target and receiving and analyzing a reflected signal from the target. A description will be given of an example where the radar is used in a vehicle. The radar may be categorized into a LRR (Long Range Radar) and an SRR (Short Range Radar) depending on the distance between the radar and the target to be detected. The LRR detects the target within a relatively long range of 100 meters, for example, and mainly utilizes an FM (Frequency Modulation) technique. On the other hand, the SRR detects the target within a relatively short range of several tens of meters, for example, and utilizes the FM technique or an SS (Spectrum Spreading) technique. The SRR is expected to be used in various applications, including control of an airbag.
FIG. 1 is a system block diagram, that is, a simulation model, of a DSSS radar. The DSSS radar first generates a PN code having a suitable length and timing by a code generator 102, and uses this PN code as a system reference signal (or baseband signal) νTX(t) which characterizes the signal transmitted from this DSSS radar. The system reference signal νTX(t) can be represented by the following formula (1), where DN denotes a device number of the DSSS radar, N denotes the length of the PN code, p(t) denotes a pulse waveform carrying the code, and Tc denotes a chip duration of the pulse. In FIG. 1, CLK denotes a clock.
                                          v            TX                    ⁡                      (            t            )                          =                              ∑                          n              =                              -                ∞                                      ∞                    ⁢                                    c                              mod                ⁡                                  (                                      n                    ,                    N                                    )                                                            D                ⁢                                                                  ⁢                N                                      ⁢                          p              ⁡                              (                                  t                  -                                      nT                    c                                                  )                                                                        (        1        )            
The system reference signal νTX(t) is up-converted by a mixer 106 using a carrier signal which is generated by a high-frequency generator (or oscillator, RF-OSC) 104 and is branched by a distributor (or hybrid circuit) 105. The up-converted transmitting signal is amplified by a high-output amplifier or a HPA (High-Power Amplifier) 108, and is transmitted from a transmitting antenna AT as a probe signal. In other words, the probe signal is obtained by modulating the carrier signal by the system reference signal νTX(t). The probe signal is transmitted by, for example, a millimeter wave signal.
In the example shown in FIG. 1, the mutual interaction between the probe signal and the target can be represented by a propagation delay time τ, a Doppler frequency fd, and an AWGN (Additive White Gaussian Noise). That is, if the target exists within an FOV (Field Of View) of the DSSS radar, the probe signal is received by the DSSS radar after receiving effects of the propagation delay time τ depending on a relative distance of the target from the radar in the direction of the line of sight (hereinafter simply referred to as a distance), the Doppler frequency fd depending on a relative velocity of the target in the direction of the line of sight (hereinafter simply referred to as a velocity), and the AWGN. In FIG. 1, reference numerals 110 and 116 denote τ/2 delays, a reference numeral 112 denotes the Doppler frequency fd, a reference numeral 114 denotes a mixer, a reference numeral 118 denotes the AWGN, and a reference numeral 120 denotes an adder, to represent the mutual interaction between the probe signal and the target in simulation.
A received signal which is received by a receiving antenna AR is amplified by an LNA (Low Noise Amplifier) 122. The received signal after this amplification is down-converted by a mixer 124 using a carrier signal which is branched by the distributor 105, and the down-converted received signal is input to a mixer 132 of a baseband demodulator 126. The mixer 132 multiplies the down-converted received signal and the PN code (or SS code for demodulation) which is delayed by a delay time τn by a delay circuit 130. An output signal of the mixer 132 is integrated by an integrator 134 for a PN code period Tf (=N*Tc), and a correlation value is output from the integrator 134. Accordingly, a baseband signal νRX(t) which is output from the mixer 132 can be represented by the following formula (2), where k (=1, . . . , K) denotes a target number specifying a k-th target of K targets, ak denotes a signal amplitude, ωdk denotes a Doppler angular frequency, τk denotes a delay time, and n(t) denotes the AWGN (Additive White Gaussian Noise).
                                          v            RX                    ⁡                      (            t            )                          =                                            ∑                              k                =                1                            K                        ⁢                                          a                k                            ⁢                              exp                ⁡                                  [                                      j                    ⁢                                                                                  ⁢                                                                  ω                        d                        k                                            ⁡                                              (                                                  t                          -                                                                                    τ                              k                                                        /                            2                                                                          )                                                                              ]                                            ⁢                                                ∑                                      m                    =                                          -                      ∞                                                        ∞                                ⁢                                                      ∑                                          n                      =                                              -                        ∞                                                              ∞                                    ⁢                                                            c                                              mod                        ⁡                                                  (                                                      m                            ,                            N                                                    )                                                                                            D                        ⁢                                                                                                  ⁢                        N                                                              ⁢                                          c                                              mod                        ⁡                                                  (                                                      n                            ,                            N                                                    )                                                                                            D                        ⁢                                                                                                  ⁢                        N                                                              ⁢                                          p                      ⁡                                              (                                                  t                          -                                                      nT                            c                                                    -                                                      τ                            k                                                                          )                                                              ×                                          p                      ⁡                                              (                                                  t                          -                                                      mT                            c                                                                          )                                                                                                                          +                      n            ⁡                          (              t              )                                                          (        2        )            
FIG. 2 is a diagram schematically showing a relationship of the SS code, the code period Tf and the chip duration Tc. In FIG. 2, the transmitting signal is illustrated as being a multiplication result of the data and the SS code, however, it is not essential for the data to be multiplied.
FIG. 3 is a diagram showing an example of a correlation output. In FIG. 3, the vertical axis indicates the correlation (the amplitude in arbitrary units), and the horizontal axis indicates the chip index. The correlation shown in FIG. 3 is obtained by plotting the output of the integrator 134 while setting the delay time τn of the delay circuit 130 to various values. The two sharp peaks in FIG. 4 indicate that at least two targets at different distances from the radar exist, and that the delay times τ of the two targets are approximately 320 chips and approximately 340 chips, respectively. A detector 136 shown in FIG. 1 detects the position (chip index) of such sharp peaks, and detection results of the detector 136 are supplied to the delay circuit 130 and a counter 138 as synchronization timings with respect to reflected signals from the two targets. In order to scan such sharp peaks for a predetermined time, the delay circuit 130 outputs the SS code to the mixer 132 while shifting the SS code by a predetermined multiple of one chip duration Tc, such as ⅓ the chip duration Tc, for example. Hence, the PN code having a predetermined delay time mTc is multiplied to the received signal in the mixer, and is integrated for the code period Tf by the integrator 134. For example, when the received code from the k-th target and the demodulated code, including the delay, become synchronized, the summation corresponding to the k-th target in the latter half (summation on the indexes m and n) of the formula (2) becomes 1, and the received signal related to this target can be represented by the following formula (2A), where a component having a synchronization error becomes
−1/N and substantially 0.νRX(t)=akexp[jωdk(t−τk/2)]+n(t)  (2A)
When the sharp peak corresponding to the target is detected, the delay circuit 130 fixes its delay time to the value which results in the sharp peak (and the synchronization timing is detected), and the synchronization is secured for the target. If a plurality of targets exist (k=1, . . . , K), the synchronization timing is detected for each target, and the above described process is carried out by the demodulator 126 in a similar manner for each of the synchronization timings. In FIG. 1, the symbol “x K” indicated above the demodulator 126 indicates that K demodulators 126 are provided in parallel, so as to facilitate the understanding of the operation. Of course, a single demodulator 126 may be used to time-divisionally carry out the above described process for each of the synchronization timings.
Next, the velocity or distance of the target from the radar is detected based on the synchronized received signal. If the velocity of light is denoted by c, a distance dk of the k-th target can be computed from the following formula. Accordingly, the resolution of the distance is proportional to the chip duration Tc.dk=(cτk)/2≈(cmTc)/2
The received signal has a Doppler frequency depending on the velocity of the target. But because the radar is a radiolocator, it is not possible to know the Doppler frequency fd in advance. In other words, it is not possible to know the velocity of the target directly from the output signal of the integrator 134. Hence, according to the conventional technique, the received signal is sampled for a duration longer than or equal to a time 1/fd (=2π/ωd=Td) which is derived from the lowest predicted Doppler frequency fd, and the sampled received signal (or reception sample) is subjected to a Fourier transform in an FFT (Fast Fourier Transform) circuit 140 in order to estimate the Doppler frequency fd. This duration Td in which the received signal is sampled becomes longer as the Doppler frequency fd becomes lower.
For example, a system for estimating the Doppler frequency using the FFT circuit in the DSSS radar has been proposed in Masahiro Watanabe et al., “A 60.5 GHz Millimeter Wave Spread Spectrum Radar and the Test Data in Several Situations”, Procedures on IEEE Intelligent Vehicle Symposium, 2002, pp. 87-91.
When the Doppler frequency fd is relatively low, the duration Td becomes relatively long. Since the velocity of the target may take various values which may be large or small, the conventional technique samples the received signal for a duration corresponding to the lowest relative velocity, for example, and estimates the Doppler frequency fd by processing the reception sample by the FFT circuit.
FIG. 4 is a diagram showing a state where a reception sample is continuously obtained for every code period Tf for each of two targets T1 and T2. In FIG. 4, the vertical axis indicates the amplitude of the Doppler signal component in arbitrary units, and the horizontal axis indicates the time (ms: milliseconds). FIG. 4 shows the reception sample after the baseband demodulation in an overlapping manner along the time base, for each of the two targets T1 and T2. Each reception sample is obtained for every integration time (Tf) of the integrator 134, and the integration time (Tf) is 0.0002 (ms) in FIG. 4. One target T1 is located at a distance of 10 meters from the radar, and is moving at a velocity of 6 km/h. The other target T2 is located at a distance of 11 meters from the radar, and is moving at a velocity of 12 km/h. In this case, it may be seen that the value of the reception sample changes at a period of approximately 0.7 (ms) for the target T2. Accordingly, if the received signal is sampled for the duration of approximately 0.7 (ms) and the reception sample is subjected to the FFT, it is possible to obtain the Doppler frequency fd and the velocity of the target T2. The received signal needs to be sampled for a duration longer than approximately 0.7 (ms) for the target T1.
Particularly in the case of radars used in vehicles, the velocity of the near target must be estimated more quickly and more accurately from the point of view of providing safety. However, problems are encountered in the conventional technique which uses the FFT in a situation where a plurality of targets are moving at various velocities. That is, in a case where the traffic is heavy on the highway and the distance between the vehicles is extremely short, for example, the time it takes to detect the velocity of the slow-moving target may govern the time required to detect the fast-moving target. Accordingly, from the point of view of realizing a high-speed and highly accurate measurement, it is undesirable to employ the technique which measures the Doppler frequency based on the FFT which requires a long time to collect the data.
Theoretically, the received signal component is obtained for every integration time NTc (=Tf) that is necessary to demodulate the PN code. Hence, there is another conventional technique which estimates the Doppler frequency by focusing on this relationship. More particularly, after the received code from the k-th target is synchronized and νRX(t)=akexp[jωdk(t−τk/2)]+n(t) is obtained for the baseband signal νRX(t) represented by the formula (2), this other conventional technique derives the Doppler frequency fd (=ωd/2π) from the value of the m-th sample of the samples that are obtained for every integration time Tf according to the following formula (3), where nR(t) denotes a real part of the AWGN n(t) and nI(t) denotes an imaginary part of the AWGN n(t).
                                          ω            d            k                    ≈                                    1                              (                                                      mT                    f                                    -                                                            τ                      k                                        /                    2                                                  )                                      ⁢                          tan                              -                1                                      ⁢                                          Im                ⁡                                  [                                                            v                      RX                      k                                        ⁡                                          (                                              mT                        f                                            )                                                        ]                                                            Re                ⁡                                  [                                                            v                      RK                      k                                        ⁡                                          (                                              mT                        f                                            )                                                        ]                                                                    ⁢                                  ⁢                                            Im              ⁡                              [                                                      v                    RX                    k                                    ⁡                                      (                                          mT                      f                                        )                                                  ]                                                    Re              ⁡                              [                                                      v                    RK                    k                                    ⁡                                      (                                          mT                      f                                        )                                                  ]                                              =                                                                                                                tan                      ⁡                                              [                                                                              ω                            d                            k                                                    ⁡                                                      (                                                                                          mT                                f                                                            -                                                                                                τ                                  k                                                                /                                2                                                                                      )                                                                          ]                                                              +                                                                                                                                                                                            n                          I                                                ⁡                                                  (                                                      mT                            f                                                    )                                                                    /                                              a                        k                                                              ⁢                                          cos                      ⁡                                              [                                                                              ω                            d                            k                                                    ⁡                                                      (                                                                                          mT                                f                                                            -                                                                                                τ                                  k                                                                /                                2                                                                                      )                                                                          ]                                                                                                                                1              +                                                                                          n                      R                                        ⁡                                          (                                              mT                        f                                            )                                                        /                                      a                    k                                                  ⁢                                  cos                  ⁡                                      [                                                                  ω                        d                        k                                            ⁡                                              (                                                                              mT                            f                                                    -                                                                                    τ                              k                                                        /                            2                                                                          )                                                              ]                                                                                                          (        3        )            
According to this other conventional technique, the Doppler angular frequency ωd can be derived quickly in a case where the SNR (Signal-to-Noise Ratio) is large. However, because the formula (3) is greatly dependent upon the noise n(t)=nR(t)+jnI(t) at a single point in time (t=mTf), there is a possibility that the accuracy of the Doppler angular frequency ωd will greatly deteriorate in a case where the SNR is poor at the point in time when the target is measured. In addition, in a case where |mTf−τk/2|→π/2ωdk, 3π/2ωdk, . . . , the accuracy of the Doppler angular frequency ωd may greatly deteriorate due to the cosine function, in the denominator, of the argument of the arctangent function approaching zero or, the tangent function, in the numerator, infinitely diverging in the positive or negative direction. The accuracy of the Doppler angular frequency ωd may also deteriorate when |mTf−τk/2|→0.